香料模拟器调音RC主动滤清器电路

通常的做法是通过将操作放大器视为理想的设备来设计主动滤清器。但是，这种简化可能导致与所需频率响应的显着偏差，这主要是由于实际放大器的有限增益带宽产品（GB）。

降低这种影响（被动/主动相补偿，预性）的常规技术不是很有效，因为它们仅考虑单杆操作模型。此外，他们忽略了其他缺陷（例如，输入，输出和负载阻抗； PIN寄生虫）。最近，加利福尼亚州圣克拉拉的国家半导体公司发表了一份设计说明¹describing an alternative and somewhat cumbersome tuning method consisting of several iteration steps with alternating calculations and measurements or simulations. The method also requires knowing all parameter sensitivities that are specific to each filter topology.

这种设计思想提出了一种更简单的技术，可以使用由现实的放大器宏观模型支持的合适的仿真软件来补偿放大器缺陷。提出的方法的基本思想是将极点频率（FP）的滤波器的循环增益调整为指定值，从而纠正极点位置，从而校正滤波器的传输函数（“ pont tuning”）。为此，必须修改反馈网络的某个部分，从而仅在一个模拟运行中计算新元素值。

所提出的技术使用“替代定理”，该技术允许线性网络的任意分支Z与独立源之间进行交换，而不会影响电压与电流比率 - 证明这些比率只有一个解决方案。²Since this is the case, specifically, for a feedback system having a loop gain of unity (A_L= 1），将人造块调谐器插入过滤器的反馈路径（请参阅图）。调谐器块的传输参数必须与倒环增益相同（幅度ALP，相φ_LP理想滤波器电路以所需频率（f）的程度（f）_Z= f_P）。For this purpose, the following expression can be used to advantage:

H_调谐器= (1/A_LP）×exp \ [ - φ_LPπs/（180 s）\] \]

where s = the complex frequency variable.

In addition, to preserve proper loading of the op amp, the TUNER input must be connected to an artificial current source simulating the output load under closedloop conditions:

I_加载= I_T/H_调谐器

where I_T=通过调谐器输出终端的电流。

提示：使用许多仿真软件包（例如MicroSIM的PSPICE）中可用的模拟行为建模技术，可以非常轻松地作为单独的子电路实现这两个函数。

In principle, the “pole-tuning” method can be applied to all second-order low-pass and bandpass topologies. The bandpass example presented in the figure is, for comparative purposes, identical to the Sallen-Key section A as given in the National Semiconductor design note. However, instead of NSC’s CLC446, a similar part will be used (the HFA1100 from Harris Semiconductor), in conjunction with a feedback resistor (R_F= 560Ω)推荐的制造商。

假设有理想的放大器，则指定的带通特性the table（第一行）可以使用此处给定的元素集实现（从NSC的设计注释中获取）：

R1 =120.3Ω\ [121 \];R4 =747.9Ω\ [750 \];R5 =20.4Ω\ [20.5 \];k = 1 + r_F/ r_G= 1.29 \ [1.294 \];r_F=560Ω\ [562 \];r_G= 1.93K \ [1.91K \];C2 = 98.67 P_F\[100 p_F\]; C3 = 10.95 p_F\[11 p_F\]

The values in brackets are the nearest nominal standard 1% value that cause, together with a realistic HFA1100 macromodel—a severe pole displacement (see the table, second row).

Setting the TUNER parameters requires the ideal loop gain of the filter to be determined first. A simulation with the ideal components and with an open feedback loop between terminals B and C yields at the pole frequency (f_P= 42.36 MHz): A_LP= 0.8937 and Φ_LP= 0.0013°.

Thus, closing the loop with a properly designed TUNER block would produce an overall loop gain A_L= 1 under ideal conditions. However, if the filter circuit contains real elements with parasitics and a realistic op-amp macromodel, the loop gain will不是统一 - 除非根据替代定理修改网络的合适部分。为此，对一个元素（在我们的情况下为R5）进行了闭环频率分析，由电压源V代替_Z(see the figure, again). Furthermore, to be realistic, standard element values are used (as given above in brackets) together with a parasitic capacitance of 0.5 pF across all resistors. Resulting from this analysis, the current-to-voltage ratio I_Z/V_Z在f_Z= f_P给出一个复杂的电导，以代替R5，可以通过两个平行参数实现：

r5* = V_Z/re（我_Z）=13.47Ω

和

C5* = IMG(I_Z）/（2πf_PV_Z）= 28.7 pf

Note that many graphical postprocessors have computational capabilities for displaying these simulation results directly with the aid of appropriate macro expressions (e.g., PSpice/Probe).

提示：作为替代方案，建议仅以一个频率（FP）运行模拟并踏入受限的C2值范围（例如，±50％）。在这种情况下，C5*和R5*可以直接显示为C2的函数，事实证明C5* = 0和R5* =16.9Ω对于C2* = 75.6 pf。因此，补偿电路瑕疵不需要额外的元素C5*。

A final simulation run using these values for C2* and R5* confirms that the dominant pole-pair of the filter circuit is shifted (i.e. “tuned”) closely to the specified position (see the table, last row).

参考：

1. Blake，K。，应用程序注释OA-29“使用调谐方法的低敏性带通滤波器设计”，《国家半导体公司》，1996年10月。
2. Desoer, C.A., and Kuh, E.S., Basic Circuit Theory, McGraw-Hill International Edition, 1969, Chapter 16.